I. INTRODUCTION

Recently, electric brake booster (EBB) systems have been widely adopted in automotive applications such as vehicles with internal combustion engines (ICEs), electric vehicles (EVs), and plug-in hybrid EVs (PHEVs).

The EBB system, which is one of the wet-type brake-by-wire systems, has replaced the mechanical brake booster not only to improve the brake performance and fuel efficiency of the vehicle but also to perform the active braking cooperative control with radar and cameras ^(1-3). It also efficiently coordinates the braking force between the regenerative braking and frictional braking in EVs and PHEVs ^(4-6).

The EBB system generally monitors the driver’s braking behavior using a brake pedal sensor and then activates the brushless direct current (BLDC) motor and solenoid valves in the EBB system. Even when the active brake systems including the anti-lock brake and electric brake control systems are not active, the EBB system must continuously operate for base braking operation. Therefore, when designing the electrical systems in automotive applications, safety considerations must be thoroughly taken into account to meet the ISO 26262 functional safety standard ⁽⁷⁾.

To comply with the above safety standard, even when the EBB system does not work properly by any failures, its operation should be promptly stopped according to the failsafe policies, and its fallback mode must be activated to proceed to the base braking operation. Fig. 1(a) and (b) show the conceptual diagrams of the EBB system that describes its operation in the normal and fallback modes, respectively. In the normal mode, when the driver puts pressure on the brake pedal, the BLDC motor is activated to provide the braking force to the wheels through the hydraulic block. Then, the two always-on solenoid valves, which are normally kept open, operate to close the hydraulic path between the 1st and 2nd chambers, thus preventing the pressure provided by the BLDC motor from leaking into the reservoir.

Fig. 1. Conceptual operation diagrams of the EBB system in the (a) normal mode, (b) fallback mode.

When a failure occurs in the EBB system, the solenoid valves including the always-on solenoid valves and BLDC motor no longer operate as the EBB system goes into the fallback mode. Although the EBB system cannot produce high pressure due to a stop in the operation of the always-on solenoid valves and BLDC motor, the piston of the 1st chamber provides the braking force to the wheels through the hydraulic block according to the pedal pressure. Even in this situation, the EBB system must still meet the minimum deceleration requirement in legislation ^(8-10).

To make the EBB system work properly, the always-on solenoid valves must operate always in the normal mode to reduce the clicking noise produced from the solenoid valve, which can be done by the pulse width modulation (PWM) ⁽¹¹⁾, while having the same feeling of operation of the existing brake system. In addition, it is necessary for the always-on solenoid valve to regulate its current to prevent hardware damages caused by thermal stress due to the heat generated by braking. Nevertheless, the solenoid valves frequently suffer from the constant current consumption that causes thermal stress. For example, a large solenoid coil current of up to 2.25 A sometimes flows to generate a desired hydraulic pressure. This leads to thermal stress that can cause a critical problem in the EBB system, resulting in a violation of the safety goal, the requirement of the ISO 26262 ASIL-D level ⁽¹²⁾.

This paper proposes a low-side solenoid valve driver that achieves a high current accuracy and dissipates less power. The proposed solenoid valve driver improves the current accuracy by controlling and averaging the solenoid coil currents, which are sensed at the moment when it is turned on and off. In addition, it reduces the power dissipation by integrating only a low-side power MOS transistor into the chip. Section II presents the architecture and operating principle of the conventional high- and low-side solenoid valve driver and the proposed low-side solenoid valve driver. In addition, a simplified four-RL branch model for solenoid coil is presented and analyzed based on the simulated and measured solenoid coil currents. Then, the proposed midpoint current sensing method using the four-RL branch model is presented and verified for its current accuracy through simulation. Section III describes the detailed circuit implementation of the proposed low-side solenoid valve driver. In Section IV, the measurement results of the fabricated solenoid valve driver are analyzed and compared with previous work. Finally, conclusions are given in Section V.

II. ARCHITECTURE AND OPERATING PRINCIPLE

1. Conventional High- and Low-side Solenoid Valve Driver

Fig. 2 shows a block diagram of the conventional high- and low-side solenoid valve driver, which is currently in production for EBB systems ^(13,14).

When the low- and high-side power MOS transistors are turned on and off, respectively, the on-state current flows through the low-side power MOS transistor, and it is sensed and mirrored by the low-side current sense-and-mirror circuit and then supplied to the analog-to-digital converter (ADC) ^(15,16). On the other hand, when the low- and high-side power MOS transistors are turned off and on, respectively, the off-state current flows through the high-side power MOS transistor, and it is sensed and mirrored by the high-side current sense-and-mirror circuit and then supplied to the ADC. These on- and off-state currents supplied to the ADC are combined and averaged, and then the averaged current is compared with a target current required to produce the necessary hydraulic pressure. The difference between the averaged current and target current is supplied to the proportional integration (PI) controller, which generates a digital output. The digital output of the PI controller is compared with the saw-tooth signal generated by the counter, and the compared result determines the duration of the PWM signal, which controls the currents flowing through the high- and low-side power MOS transistors.

Fig. 2. Block diagram of the conventional high- and low-side solenoid valve driver.

In this way, the conventional high- and low-side solenoid valve driver ⁽¹³⁾ senses the on- and off-state currents. However, since the high-side power MOS transistor is located inside the IC, the off-state current flows into the IC, resulting in a large power consumption. Moreover, since the EBB system requires solenoid valves with more than 14-channels ⁽¹⁷⁾, it can suffer from thermal stress due to excessive power dissipation if all the solenoid valves are implemented with the conventional high- and low-side solenoid valve driver. Therefore, it is necessary to avoid the off-state current flowing into the IC, which has been a power dissipation problem in the conventional high- and low-side solenoid valve driver.

2. Proposed Solenoid Valve Driver

Fig. 3 shows the proposed low-side solenoid valve driver with an external freewheeling diode, which eliminates the necessity of the high-side current sense-and-mirror circuit and high-side power MOS transistor used in the conventional solenoid driver.

Fig. 3. Block diagram of the proposed low-side solenoid valve driver.

In the proposed low-side solenoid valve driver, the solenoid coil current is sensed and mirrored using a low-side current sense-and-mirror circuit. The sensed current is then sampled and digitized through an ADC at the moment when the low-side power MOS transistor is turned on and off. Next, the digitized current is averaged by a digital filter, and then the averaged current is compared with a target current required to produce the necessary hydraulic pressure. The difference between the averaged current and target current is supplied to the PI controller. The digital output of the PI controller is compared with the saw-tooth signal generated by the counter, and the compared result determines the duration of the PWM signal, which controls the on-state current flowing through the low-side power MOS transistor.

Assuming that the battery voltage ($V$) is 14 V and the target current (${I_{TARGET}}$) is 2 A, the power dissipation of both the drivers ($P$) can be expressed as in ⁽¹³⁾

(1)

$P=R_{ON\_ LS}I_{ON}^{2}\alpha +R_{ON\_ HS}I_{OFF}^{2}(1- \alpha )+VI_{ON}t_{SW}f_{SW}$,

where ${R_{ON\_LS}}$ and ${R_{ON\_HS}}$ are the on-resistances of the low- and high-side power MOS transistors, ${I_{ON}}$ and ${I_{OFF}}$ are the on- and off-state currents, respectively, ${\alpha}$ is a duty cycle of the PWM signal, ${t_{SW}}$ is the switching time, and ${f_{SW}}$ is the switching frequency.

Table 1 shows the design parameters of the conventional and proposed solenoid valve drivers, indicating that owing to the exclusion of the power consumption caused by ${R_{ON\_HS}}$ in the conventional high- and low-side valve driver, the power consumption of the proposed solenoid valve driver was reduced by 44% compared with that of the conventional one under the same design conditions.

Table 1. Design parameters of the solenoid valve drivers

Design parameters	Unit	Conventional high- and low-side driver	Proposed low-side driver
$V$	volts	14	14
$I_{TARGET}$	amperes	2	2
$f_{SW}$	kHz	4	4
$R_{ON_HS}$	ohms	0.25	-
$R_{ON\_LS}$	ohms	0.25	0.25
α	%	50%	50%
$t_{SW}$	μs	1.2	1.2
P	watts	1.1344	0.6344

3. Proposed Solenoid Coil Model and Midpoint Current Sensing Method

Unlike the conventional high- and low-side solenoid valve driver, the proposed low-side solenoid valve driver employs the external freewheeling diode to protect the low-side power MOS transistor by discharging energy in the inductor through the external freewheeling diode, resulting in a less power dissipation, less thermal stress, and lower design complexity. However, since the low-side solenoid valve driver senses the on-state current only, it suffers from poor current accuracy ⁽¹³⁾. To improve current accuracy, the Ton/2 current sensing method, which senses the solenoid coil current at half the duration of the on-state current (Ton), has been studied for solenoid valve drivers ^(18,19). However, this current sensing method inaccurately senses the current flowing through the solenoid coil, thus producing a large current difference between the measured and sensed solenoid coil currents. Fig. 4 shows the measured solenoid coil current with respect to the measured drain voltage of the low-side power MOS transistor at a PWM duty cycle of 10%. Here, the average current measured using an oscilloscope is 270 mA, while the sensed current using the Ton/2 current sensing method is 340 mA, resulting in a current difference of 70 mA, indicating that the conventional Ton/2 current sensing method is not accurate. Such a large current difference is mainly caused by the variation in the impedance value of the solenoid valve, which occurs during the solenoid valve operation ^(20-23), and thereby the conventional Ton/2 current sensing method is not suitable for solenoid valve drivers. Therefore, a sensing method considering the above variation in the impedance of the solenoid valve is necessary to accurately sense the solenoid coil current.

Fig. 4. Measured solenoid coil currents with respect to the measured drain voltage at a PWM duty cycle of 10%.

Fig. 5. (a) Current waveform of the solenoid coil with different slopes according to the time, (b) the four-RL branch model representing the different slopes.

To analyze the above current difference due to the impedance variation, the solenoid coil current waveform is investigated as follows. Fig. 5(a) and (b) respectively show the solenoid coil current waveform with different slopes according to the time and the four-RL branch model employed to represent the different approximate slopes of the current waveform ^(22,23). The transfer function of the four-RL branch model, ${h(t)}$, which is used to model the inverse of the impedance of the solenoid valve, can be expressed as in ⁽²⁴⁾

(2)

\begin{equation} h\left(t\right)=\sum _{i=1}^{4}\frac{1}{R_{i}}\left(1- e^{\frac{R_{i}}{L_{i}}t}\right)u\left(t\right), \end{equation}

where ${R}_{i}$ is the i$^{\mathrm{th}}$ resistance, ${L}_{i}$ is the i$^{\mathrm{th}}$ inductance, ${t}$ is the time, and ${u(t)}$ is a unit step function. By multiplying the battery voltage ($V_{BAT}$) by ${h(t)}$in (2), the highest and lowest solenoid coil currents (${I}_{HIGH}$ and ${I}_{LOW}$) considering the duty cycle of the PWM signal ($\alpha $) can be respectively derived as

Fig. 6. Measured solenoid coil current and the simulated current using the four-RL branch model at a target current of 1 A when the battery voltage is (a) 9 V, (b) 16 V.

(3)

$I_{HIGH}=\sum _{i=1}^{4}\frac{V_{BAT}}{R_{i}}\left(\frac{1- e^{\frac{R_{i}}{L_{i}}\alpha T}}{1- e^{\frac{R_{i}}{L_{i}}T}}\right),$

(4)

\begin{equation} I_{LOW}=\sum _{i=1}^{4}\frac{V_{BAT}}{R_{i}}\left(\frac{\left(e^{\frac{R_{i}}{L_{i}}\alpha T}- 1\right)e^{\frac{R_{i}}{L_{i}}T}}{1- e^{\frac{R_{i}}{L_{i}}T}}\right), \end{equation}

where ${T}$ is the period of the PWM signal, ^(21,24). Here, ${I}_{LOW}$ and ${I}_{HIGH}$ represent the currents at the moment when the low-side power MOS transistor is turned on and off, respectively.

Using (2), (3), and (4), an iterative simulation is carried out to fit the four-RL branch model to the measured current waveforms by adjusting the values of ${R}$ and ${L}$. Fig. 6(a) and (b) show the measured solenoid coil current and the simulated current using the four-RL branch model at a target current (${I_{TARGET}}$) of 1 A when the battery voltage is 9 V and 16 V, respectively. Here, the measured and simulated currents are depicted in red and blue, respectively, showing a good match between the two currents. Therefore, the four- RL branch model can be applied to the new current sensing method to obtain an accurate solenoid coil current approximating the measured current.

Fig. 7 shows the simulated solenoid coil current using the four-RL branch model and its average current at a PWM duty cycle of 50%. Since the area of the simulated current waveform using the four-RL branch model is equal to that of the dotted triangle, the average value of the simulated current can be obtained from half the peak- to-peak value of the simulated current, which is equal to half the ${I}_{LOW}$-to-${I}_{HIGH}$ value, representing its midpoint. As shown in Fig. 7, the midpoint value of the simulated current using the four-RL branch model and the measured solenoid coil current are almost equal to a current value of 2.1 A, while the simulated current using the conventional Ton/2 current sensing method has a difference of 65 mA compared to the measured solenoid coil current. Consequently, the solenoid coil current can be accurately obtained from the midpoint of the currents at the moment when the low-side power MOS transistor is turned on and off. For the reason described above, in this work, such a midpoint current sensing method using the four-RL branch model is employed to improve the accuracy of the solenoid coil current.

Fig. 7. Simulated current using the four-RL branch model and its average current, and the conventional Ton/2 current at a PWM duty cycle of 50%.

Fig. 8. Simulation results for difference between the target and sensed currents using both the sensing methods.

To further compare the performance of the proposed midpoint sensing method and the conventional Ton/2 current sensing method, the simulation for the difference between the target and sensed currents using both the sensing methods is carried out with respect to the target current, as shown in Fig. 8. The simulation results show that the proposed midpoint sensing method has a much smaller difference than the conventional Ton/2 sensing method, demonstrating that the proposed midpoint sensing method senses the solenoid coil current more accurately than the conventional Ton/2 current sensing method.

Fig. 9. Detailed low-side current sense-and-mirror circuit and ADC in the proposed solenoid valve driver.

III. CIRCUIT DESCRIPTION

Fig. 9 shows the detailed low-side current sense-and-mirror circuit and ADC in the proposed low-side solenoid valve driver.

When the voltage at node ${N}_{1}$ of ${M}_{1}$ ($V_{1}$) is high, transistors, ${M}_{1}$${-M}_{5}$, are turned on, where ${M}_{1}$ and ${M}_{2}$ operate in the deep triode region, while ${M}_{3}$, ${M}_{4}$, and ${M}_{5}$ operate in the saturation region. Then, the current sense amplifier (CSA) in the low-side current sense-and-mirror circuit senses the current flowing through ${M}_{1}$ (${I}_{LOAD}$) and produces a voltage at node ${N}_{3}$, which is equal to the voltage at node ${N}_{2}$. Therefore, the mirrored current (${I}_{MIRROR}$), which has one-m$^{\mathrm{th}}$ of ${I}_{LOAD}$, flows through ${M}_{2}$${-M}_{5}$ into the input of the current comparator ($CC_{SIGN}$) to compare ${I}_{MIRROR}$ with a current generated from the DAC (${I}_{DAC}$), where m is the transistor size ratio of ${M}_{1}$/${M}_{2}$.

When ${I}_{MIRROR}$ is less than ${I}_{DAC}$, the voltage at node ${N}_{3}$ decreases and turns off ${M}_{C4}$, and thus the ${SIGN}$bit remains high and the digital logic increases ${I}_{DAC}$ to digitize ${I}_{MIRROR}$ properly in the next phase.

When ${I}_{MIRROR}$ is greater than ${I}_{DAC}$, the voltage at node ${N}_{3}$ increases and turns on ${M}_{C4}$, thus pulling down the voltage at node ${N}_{4}$ and making the ${SIGN}$bit low. Subsequently, ${M}_{C2}$ is turned on, and thus ${I}_{ERR}$ flowing through ${M}_{C2}$ and ${M}_{C3}$ is mirrored to the input of ${M}_{C5}$ and ${M}_{C6}$ in the current comparator ($CC_{ERR}$) consisting of 10 comparator cells, each of which has a pull-down current. The $CC_{ERR}$ compares ${I}_{ERR}$ with the corresponding binary-weighted ${I}_{REF}$ generated from the bandgap reference voltage generator and ${M}_{G1}$, and then produces an ${ERR}$ bit. Then, the digital logic decreases ${I}_{DAC}$ to digitize ${I}_{MIRROR}$ properly in the next phase.

Fig. 10. Simulation results of the proposed low-side solenoid valve driver when the target current is 1 A at a PWM frequency of 4 kHz.

In this way, ${I}_{MIRROR}$ is compared with ${I}_{DAC}$ continuously and digitized into the ${SIGN}$ and ${ERR}$ bits at the moment when ${M}_{1}$ is turned on and off by the digital logic. Then, the digitized bits of ${I}_{MIRROR}$ are transmitted to the digital filter shown in Fig. 3 to obtain the midpoint current by averaging these digitized bits.

When $V_{1}$ is low, ${M}_{1}$ is turned off and the solenoid coil current is recirculated through the freewheeling diode.

Fig. 10 shows the simulation results of the proposed low-side solenoid valve driver when the target current is 1 A at a PWM frequency of 4 kHz, representing ${I}_{LOAD}$ reaches a target current of 1 A when the voltage at node ${N}_{2}$ ($V_{2}$) is 14 V. Once ${I}_{LOAD}$ reaches a target current of 1~A, the proposed low-side solenoid valve driver regulates ${I}_{LOAD}$ to maintain the target current.

Fig. 11. Thermal simulation results of the high- and low-side solenoid valve driver and the proposed low-side solenoid valve driver.

To verify the robustness of the proposed solenoid valve driver under the worst-case conditions in the system level, the worst-case analysis is carried out with an ${R}_{ds,on}$ value of 250 mΩ and an ${M}_{1}$ size of 0.51 mm$^{2}$. Here, ${R}_{ds,on}$ is the on-resistance in the drain-to-source of the transistor. In the case of the always-on solenoid valve, the ${R}_{ds,on}$ value was determined to be as low as possible to reduce the power dissipation because the solenoid valve must continuously operate even when the vehicle is stopped by braking operation.

Fig. 11 shows the thermal simulation results of the conventional high- and low-side solenoid valve driver ⁽¹³⁾ and the proposed low-side solenoid valve driver at an ${R}_{ds,on}$ value of 250 mΩ when all the solenoid valve drivers operate for 1,000 seconds. The environment for thermal simulation is configured under the following boundary conditions. The heat of the IC is dissipated using the external heat sink and metal cover. The printed-circuit-board (PCB) is composed of 6-layers with thermal VIAs having a filling ratio of 50%. The heat sink copper block is soldered at the bottom of the PCB and connected to the metal cover with the insulator to prevent it from being shorted. As shown in Fig. 11, the die temperature, which is a delta junction temperature, rises sharply as the current flows through the power MOS transistor. After a certain time has elapsed, it becomes saturated owing to the external heat sink and metal cover. The junction temperature of the proposed low-side solenoid valve driver is saturated to 164 $^{\circ}$C at an engine room temperature of 125 $^{\circ}$C, which is about 5 $^{\circ}$C lower than that of the high- and low-side solenoid valve driver.

Fig. 13. Measured $V_{BAT}$, $V_{1}$, $V_{2}$, and $I_{LOAD}$ of the proposed low-side solenoid valve driver when the target current is 0.4 A at a $V_{BAT}$ of 14 V.

To meet the requirement of a junction temperature of 175 $^{\circ}$C, the BCDMOS process technology was used in this work.

IV. MEASUREMENT RESULTS

Fig. 12 shows a photomicrograph of the proposed low-side solenoid valve driver, which was fabricated using a 0.11 μm BCDMOS process. The fabricated IC was measured using a solenoid valve with the freewheeling diode. Fig. 13 shows measured battery voltage ($V_{BAT}$), $V_{1}$, $V_{2}$, and ${I}_{LOAD}$ when ${I_{TARGET}}$ is 0.4 A at a battery voltage of 14 V. $V_{1}$ was measured to be 3.3 V in a high state and 0~V in a low state. When $V_{1}$ is low or high, the measured $V_{2}$ is respectively equal to a battery voltage of 14 V or about 100 mV, which is obtained by multiplying ${Rds,on}$of ${M}_{1}$ (250 mΩ) and an ${I}_{LOAD}$ of 400 mA, demonstrating that the proposed low-side solenoid valve driver works properly.

Fig. 14. Measurement results for the difference between the sensed and target currents with respect to the target current using both the proposed midpoint and conventional sensing methods at different temperatures of -40 °C, 25 °C, and 125 °C when the battery voltage is 14 V.

Fig. 14 shows the measurement results for the difference between the sensed and target currents with respect to the target current using both the proposed and conventional sensing methods at different temperatures of -40 $^{\circ}$C, 25 $^{\circ}$C, and 125 $^{\circ}$C when the battery voltage is 14 V. These measurement results show a tendency similar to the simulation results shown in Fig. 8, demonstrating that the proposed midpoint current sensing method achieves a much smaller difference than the conventional Ton/2 current sensing method.

Table 2 shows the performance comparison of the proposed solenoid valve driver with previous works.

As the target current increases, the current sensing accuracy of the proposed midpoint sensing method becomes significantly better than that of the conventional Ton/2 sensing method, showing a much better current accuracy especially in the region of a target current higher than 0.5 A, where the EBB system mainly operates.

V. CONCLUSIONS

In this paper, we propose a highly accurate low-side solenoid valve driver using the midpoint current sensing method for EBB systems. The proposed midpoint current sensing method senses the solenoid coil current at the moments when the low-side solenoid valve driver is turned on and off, and then obtains the peak-to-peak value of the sensed currents, thus improving the current accuracy of the solenoid coil. In addition, the proposed low-side solenoid valve driver integrates only a low-side power MOS transistor into the chip to avoid the off-state current flowing into the IC, which has been a power dissipation problem in the conventional high- and low-side solenoid valve driver, thus operating with a power dissipation that is 44% less than that of the conventional solenoid valve driver. The measurement shows that the current accuracy is improved by up to 8% compared with the conventional Ton/2 current sensing method. Therefore, the proposed low-side solenoid valve driver is suitable for EBB systems requiring highly accurate current and low power consumption.